The present invention relates to an apparatus and a method for sensing a current within a coil, said current being applied to the coil such that it is switched via a switching transistor.
In a system, a current value may be sensed, for example, for detecting an overload operation or a short-circuit. However, the current value may also be used as a controlled variable, such as in regulated current sources. Potential fields of application of the invention are any kinds of voltage and current transformers, in particular CMOS-integrated DC/DC converters.
So that the design volume of the passive devices in the power supply part in mobile systems may be further reduced, it may be useful, for example, to further increase the switching frequency employed in DC/DC converters. In such voltage supply parts it is important to monitor the values of the currents flowing in the various circuit parts, for example to detect the system being overloaded or to check a regulated current source for charging a lithium ion accumulator. In these applications, what is particularly important is the average value of the current flowing in the power path, viewed over a clock period.
To perform a current measurement, the current value is often first of all converted to a proportional voltage value, e.g. using a resistor or a Hall sensor, since voltages are easier to evaluate, for example by means of an analog/digital converter (ADC). At high clock frequencies, measuring currents becomes particularly problematic. Current sensors comprising operational amplifiers having large bandwidths may be used. Such operational amplifiers consume a considerable amount of quiescent current, which negatively affects the efficiency factor of the entire DC/DC converter. If, in addition, the average value of the current may be used for the application, it may be useful to either calculate this value in a digital manner or to integrate it in an analog manner. This results in the requirement of a costly and complex circuit, which will also consume a large amount of energy.
The requirements placed on a current sensor applied in mobile systems typically comprise low energy absorption and high accuracy. If the current sensor is to operate even at high switching frequencies, the signal/noise ratio may become problematic, since relatively steep switching edges generate electromagnetic interference. If the current sensor is employed in power converters for battery-operated mobile devices, it should be insensitive to temperature changes and ensure high accuracy over the entire battery voltage range. To combine said requirements within one sensor represents a major challenge.
This problem is to be discussed in further detail by means of an example: for modern mobile devices, lithium ion batteries are mostly employed, which provide a nominal voltage of about 3.6 V. When the battery cell is fully charged, it even has a voltage of 4.2 V. The period duration of a 10 MHz DC/DC converter is 100 ns, of which about 1/10 of the period is used up by the switching transients. This means that the rising and falling edges each have a maximum duration of about 10 ns. If the switched voltage of 4.2 V is to be set at the operational amplifier, this will correspond to a slew rate of 420 V/μs (4.2 V/10 ns). The current consumption of such an operational amplifier ranges from 5 to 10 mA. To ensure low signal distortion, the frequency bandwidth should at least range from 50 to 100 MHz. A mobile telephone has a quiescent current consumption in the range from 3 to 6 mA, for example. If the current sensor causes an additional current consumption of 5-10 mA, this represents a duplication of the overall current consumption. This results in that about half up to three quarters of the battery energy would be consumed within the current sensor, which would be extremely unsuitable for a mobile application. Therefore, in this context, a current sensor would be desirable which enable measuring the average value of the current without requiring devices having high bandwidths, i.e. having high current consumptions at the same time.
In mobile systems, sampling of currents which are switched at a high frequency represents a particular challenge, since the energy consumption of the current sensor should be kept as low as possible. FIGS. 1a-1d depict the current measurement variants currently complying with conventional technology which are employed in integrated DC/DC converters. As an example, a buck-converter topology was selected which has a power switch M1, a power diode D1, and a power coil L1. The input voltage Vin is also the supply voltage, which in mobile devices corresponds to the battery voltage and is converted to the output voltage Vout. A capacitor Cout smoothens the output voltage Vout.
In FIG. 1a, a power switch M1 is connected between a voltage input 10 and a branching node 12. The power switch M1 comprises a control input 14 to which a control signal may be applied in order to switch it. A coil L1 and a resistor Rs are connected in series between the branching node 12 and a voltage output 16. A power diode D1 is reverse-connected between the branching node 12 and a reference node 18. A capacitor Cout is connected between the voltage output 16 and the reference node 18. An input voltage Vin, or a supply voltage, is present at the voltage input 10. A current ID(M1) flows through the power switch M1. An output voltage Vout is available at the voltage output 16. The voltage Vsense may be measured across the resistor Rs.
The current sensor in FIG. 1a comprises a resistor Rs across which the voltage Vsense is measured. As is described in Ohm's law, the measured voltage Vsense is proportional to the current value IL1. The proportionality constant is the selected resistance, which is well defined. This current sensor enables measuring the present current value ID(M1) within the power switch M1 with high accuracy. However, this method has many disadvantages. Since the resistor Rs is located within the current path, it causes line losses which increase as a square of the flowing current IL1.
In FIG. 1b, a power switch M1 is connected between a voltage input 20 and a branching node 22. A coil L1 is connected between the branching node 22 and a voltage output 26. A power diode D1 is reverse-connected between the branching node 22 and a reference node 28. A capacitor Cout is connected between the voltage output 26 and the reference node 28. An input voltage Vin is present at the voltage input 20, an output voltage Vout is available at the voltage output 26. The power switch M1 comprises a control input 24 to which a control signal may be applied so as to switch the power switch M1. A current ID(M1) flows through the power switch M1. A current IL1 flows through the coil L1. A voltage drop Vsense across the power switch M1 may be measured.
The current sensor in FIG. 1b samples the voltage drop Vsense across the power switch M1. This method causes no additional line losses within the current path, but the accuracy that may be achieved is substantially lower, since the voltage drop Vsense is highly dependent on the temperature and on the drive. Due to the fact that the power switch M1 is switched, the current flowing through it will undergo intense dynamic changes. Delay times are to be taken into account for the current sensor so that the current measured has achieved a stable value.
In FIG. 1c, a power switch M1 is connected between a voltage input 30 and a branching node 32. The power switch M1 comprises a control input 34 to which a control signal may be applied to switch the power switch M1. A coil L1 is connected between the branching node 32 and a voltage output 36. A power diode D1 is reverse-connected between the branching node 32 and a reference node 38. A capacitor Cout is connected between the voltage output 36 and the reference node 38. In parallel with the coil L1, a second capacitor Cs and a resistor Rs are connected in series between the branching node 32 and the voltage output 36. An input voltage Vin is present at the voltage input 30, an output voltage Vout is available at the voltage output 36. A current ID(M1) flows through the power switch M1. A current IL1 flows through the coil L1. A voltage Vsense may be measured across the second capacitor Cs.
The current sensor in FIG. 1c samples the voltage Vsense at the coil L1 via a low-pass filter (Rs, Cs). The voltage Vsense at the output of the low-pass filter is an image of the current IL1 through the coil L1. This method causes hardly any additional line losses in the current path. However, the low-pass filter (Rs, Cs) is to be precisely adapted to the coil L1. To this end, highly precise passive devices (Rs, Cs) are to be employed which mostly may only be employed in a discrete design. This means that this method is not suitable for a monolithically integrated DC/DC converter.
In FIG. 1d, a power switch M1 is connected between a voltage input 40 and a branching node 42. The power switch M1 comprises a control input 44 to which a control signal may be applied to switch the power switch M1. A coil L1 is connected between the branching node 42 and a voltage output 46. A power diode D1 is reverse-connected between the branching node 42 and a reference node 48. A capacitor Cout is connected between the voltage output 46 and the reference node 48. A sensor switch M1a is connected between the voltage input 40 and a sensor output 50. The sensor switch M1a comprises a sensor control input 52 which is interconnected to the control input 44 of the power switch M1. A current ID(M1) flows through the power switch M1. A current IL1 flows through the coil L1, and a current Isense flows through the sensor switch M1a.
The current sensor in FIG. 1d samples the current ID(M1) over a current mirror (M1, M1a). The current mirror (M1, M1a) generates a proportional current Isense which is smaller, by a factor of about 10 to 10,000, than the current ID(M1) to be measured. This method may easily be implemented in CMOS circuits. However, it offers only a low level of accuracy if the two switches (M1, M1a) within the current mirror do not have the same working point.
To remedy the problem of the lack of accuracy of the current sensor in FIG. 1d, it is customary to match the working points of the two switches (M1, M1a) within the current mirror. This principle is shown in FIG. 2.
In a conventional current sensor, shown in FIG. 2, for example, the voltage VSW1 at the power coil L1 is connected between ground VSSA and supply voltage Vin. If a 10 MHz pulse width modulation (PWM) signal is employed, for example, to drive the control input 64 of the power switch M1 with a control signal VG(M1), the operational amplifier 74 should have a very high bandwidth. This means high energy consumption, which restricts application in mobile devices, since this energy is also consumed in the standby mode of the mobile device. Therefore, the operating period would be highly reduced.
FIG. 2 shows the conventional current sensor 60, which matches the working points of a power switch M1 and of a sensor switch M1a. The conventional current sensor 60 comprises a sensor switch M1a, an operational amplifier 74, a measuring transistor M1b and a sensor resistor Rsense. A power switch M1 is connected between a voltage input 61 and a branching node 62. The power switch M1 comprises a control input 64 to drive the power switch M1 by means of a control signal VG(M1). A power diode D1 is reverse-connected between the branching node 62 and a reference node 68. In parallel with the power diode D1, a series connection of a (power) coil L1 and an output capacitor Cout is connected between the branching node 62 and the reference node 68. An output voltage Vout is available between the coil L1 and the output capacitor Cout. The sensor switch M1a is switched such that an input of the sensor switch M1a is interconnected to an input of the power switch M1, both inputs being connected to the voltage input 61. The sensor switch M1a comprises a sensor control input 72 connected to the control input 64 of the power switch M1. The control signal VG(M1) is present at the sensor control input 72 and at the control input 64.
The operational amplifier 74 comprises a negative input connected to an output of the sensor switch M1a, and a positive input connected to the branching node 62. The measuring transistor M1b comprises a measuring transistor control input 76 connected to an output of the operational amplifier 74. An input of the measuring transistor M1b is connected to the output of the sensor switch M1a and to the negative input of the operational amplifier 74. An output of the measuring transistor M1b is connected to the reference node 68 via a sensor resistor Rsense.
An input voltage Vin is present at the voltage input 61. A current ID(M1) flows through the power switch M1. A sensor current Isense flows through the sensor switch M1a. A coil current IL1 flows through the coil L1. A voltage VSW1 drops at the branching node 62. A sensor voltage Vsense drops at the sensor resistor Rsense.
The current Isense, which is an image of the current ID(M1), generates the voltage Vsense by means of the resistor Rsense. An operational amplifier 74 having a high bandwidth is employed to minimize the voltage difference between the two switches of the current mirror. If high clock frequencies are employed for the power switch M1, the operational amplifier 74 should offer a large bandwidth. Operational amplifiers having large bandwidths consume a large amount of energy, however, which renders them unsuitable for being employed in mobile systems.
A typical DC/DC voltage converter, for which a full-bridge topology is employed, is depicted in FIG. 3. FIG. 3 shows a circuit diagram of a conventional full-bridge circuit in a buck mode topology. The full-bridge circuit 100 comprises a power input circuit 101, an H bridge circuit 102, which is configured in buck mode topology, a power output circuit 103, and a pulse width modulation control circuit 104 which is operated in a voltage mode.
The power input circuit 101 comprises a voltage source 108 connected between a reference node 109 and an output 110 of the power input circuit 101. An input voltage Vin is generated at the voltage source 108, and an input current Iin is provided at the output 110.
The H bridge circuit 102 comprises four power switches M1, M2, M3, M4, a coil L1, and an output capacitor Cout. A first input 111 of the H bridge circuit 102 is interconnected to the output 110 of the power input circuit 101. The H bridge circuit 102 comprises a second input 112 and an output 113.
The first power switch M1 is connected between the first input 111 and a first branching node 116 of the H bridge circuit 102. The first power switch M1 comprises a first control input 120 interconnected with the second input 112 of the H bridge circuit 102. The second power switch M2 is connected between the first branching node 116 and a reference node 109 of the H bridge circuit 102. The second power switch M2 comprises a control input 122 connected to the reference node 109. The coil L1 is connected between the first branching node 116 and a second branching node 118. The third power switch M3 is connected between the second branching node 118 and the reference node 109. The third power switch M3 comprises a control input 124 connected to the reference node 109. The fourth power switch M4 is connected between the second branching node 118 and the output 113 of the H bridge circuit 102 and comprises a control input 126 connected to the reference node 109. The output capacitor Cout is connected between the output 113 of the H bridge circuit 102 and the reference node 109. A current IL1 flows through the coil L1. The control input 120 of the first power switch M1 receives a control signal PWM BUCK.
The power output circuit 103 comprises an output impedance Zout connected between an input 130 of the power output circuit 103 and a reference node 109 of the power output circuit 103. The input 130 of the power output circuit 103 is connected to the output 113 of the H bridge circuit 102. The input 130 of the power output circuit 103 receives an output current Iout flowing through the output impedance Zout, so that an output voltage Vout drops at the output impedance Zout.
The pulse width modulation control circuit 104 comprises a reference voltage source 140, an operational amplifier 142, a comparator 144, a first feedback resistor RFB1 and a second feedback resistor RFB2. The pulse width modulation control circuit 104 comprises an input 146 connected to the output 113 of the H bridge circuit 102 and to the input 130 of the power output circuit 103. The pulse width modulation control circuit 104 comprises an output 148 connected to the second input 112 of the H bridge circuit 102.
The reference voltage source 140 is connected between a positive input of the first operational amplifier 142 and a reference node 109 of the pulse width modulation control circuit 104. The reference voltage source 140 generates a reference voltage Vref. A negative input of the operational amplifier 142 is connected to a branching node 150 of the pulse width modulation control circuit 104. An output of the operational amplifier 142 is connected to a negative input of the comparator 144. A fault voltage VEA is present at the negative input of the comparator 144, an oscillator voltage VOSC is present at a positive input of the comparator 144. The output of the comparator 144 is connected to the output 148 of the pulse width modulation control circuit 104 and provides the control signal PWM BUCK. The first feedback resistor RFB1 is connected between the branching node 150 of the pulse width modulation control circuit 104 and the reference node 109. The second feedback resistor RFB2 is connected between the input 146 and the branching node 150 of the pulse width modulation control circuit 104. A feedback voltage VFB drops at the first feedback resistor RFB1. The reference nodes 109 of the four circuits 101, 102, 103, 104 are at ground potential.
A full bridge comprises a discrete power coil L1 connected to four power switches (M1, M2, M3, M4). The impedance of the load Zout is connected to the output voltage Vout. The output voltage Vout is adjusted by a voltage divider which is based on resistors (RFB1, RFB2) and which generates the feedback voltage VFB. A fault amplifier 142 amplifies the voltage difference between the feedback voltage VFB and the reference voltage Vref and thus generates a fault voltage VEA. A pulse width modulator (PWM modulator) 104 generates the PWM pulses (PWM BUCK) on the basis of the fault voltage VEA and of the carrier signal Vosc.
The full-bridge topology enables realizing the buck converter, boost converter and buck-boost converter topologies.
At high clock frequencies, which may be used in the full-bridge circuit, utilization of a conventional current sensor of FIG. 2 in the full-bridge circuit of FIG. 3 may render precise measurement of currents extremely expensive. For precise measurement of the currents, operational amplifiers having high bandwidths may be used which consume a large amount of quiescent current and may therefore negatively affect the efficiency factor of the entire bridge circuit. If the average value of the current may be used, a further circuit may be used for determining this value, which circuit will then consume further energy. At high switching frequencies, the signal/noise ratio may deteriorate due to the relatively steep switching edges and to the electromagnetic interference effects associated therewith, and may impair the accuracy of the current measurement.